Multiple trellis coded differential unitary space-time modulation

ABSTRACT

A method of differential unitary space-time modulation is provided by encoding messages to transmit through multiple transmitting antenna into multiple trellis code in a communication system wherein the signals are transmitted between multiple transmitting antennas and at least one receive antenna. The performance provided is superior to that of the modulation of the prior art.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a method of differential unitary space-time modulation by encoding messages to transmit through multiple transmit antenna into multiple trellis code in a communication system wherein the signals are transmitted between multiple transmit antennas and at least one receive antenna.

2. Prior Art

Differential unitary space-time modulation (DUSTM) is a scheme proposed for the transmit antenna diversity to combat detrimental effects in wireless fading channels at a receiver without channel state information. In the DUSTM, the transmitted signal matrix at each time block is the product of the previously transmitted signal matrix and the current unitary data matrix. The constellations for unitary space-time modulated signals form groups under matrix multiplication.

Since the proposal of the DUSTM, several works to improve performance of the DUSTM have been introduced.

However, the performance of DUSTM in a communication system wherein the signals are transmitted and received between multiple transmit antennas and at least one receive antenna is not satisfactory.

SUMMARY OF THE INVENTION

One object of the invention is to provide a method of DUSTM by encoding messages to transmit through multiple transmitting antenna into multiple trellis code in a communication system wherein the signals are transmitted between multiple transmit antennas and receive antenna.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a transmitter used in the DUSTM according to the present invention.

FIG. 2 illustrates a set partitioning for (8; [1,3]) in the DUSTM according to the present invention.

FIGS. 3 a and 3 b illustrate the multiple trellis code wherein the rate is 4/6, and the number of states are 4 and 8, respectively.

FIGS. 4 a and 4 b illustrate bit error rate (BER) in the multiple trellis coded (MTC) DUSTM, uncoded DUSTM, and trellis coded DUSTM, wherein R is 1 bit/s/Hz and 1.5 bits/s/Hz, respectively.

FIG. 5 is a flow chart illustrating the method of DUSTM according to the present invention.

DETAILED DESCRIPTION OF THE PREFERRED IMPLEMENTATION

Reference will now be made in detail to the modulation method according to preferred embodiments of the present invention as illustrated in the accompanying drawings.

First, the channel model will be explained. It is assumed that the channel is a wireless channel in which data are sent from n_(T) transmit antennas to n_(R) receive antennas. n_(T) is an integer no less than 2, and n_(R) is an integer no less than 1.

Further, it is assumed that this channel has flat Rayleigh fading, and the channel coefficients for different transmit-receive antenna pairs are statistically independent and remain unchanged during T time interval of symbolic period. Let c_(t) ^(j) denote the transmitted signal from the transmit antenna j at the time t where j=1, 2, . . . , n_(T), and t=1, 2, . . . , T. The received signal y_(t) ^(j) for the received antenna i at the time t is given by Formula 1:

$\begin{matrix} {{y_{t}^{i} = {{\sqrt{\rho}{\sum\limits_{j = 1}^{n_{T}}{h_{ji}c_{t}^{j}}}} + n_{t}^{i}}},{i = 1},2,\ldots\mspace{14mu},n_{R}} & \left\lbrack {{Formula}\mspace{14mu} 1} \right\rbrack \end{matrix}$

wherein,

ρ represents the signal-to-noise ratio (SNR) per receive antenna,

h_(ji) represents the zero-mean complex Gaussian channel coefficient from the transmit antenna j to the receive antenna i with variance 0.5 per dimension, and

n_(t) ^(i) is the zero-mean complex additive white Gaussian noise (AWGN) at the receive antenna i with variance 0.5 per dimension.

It is assumed that the power of transmitted signal is normalized to unit power.

The received signal in matrix form, i.e., received signal matrix, is written as Formula 2: Y=√{square root over (ρ)}CH+N  [Formula 2]

wherein,

Y={Y_(t) ^(i)} is the T×n_(R) received signal matrix,

C={C_(t) ^(j)} is the T×n_(T) transmitted signal matrix,

H={h_(ji)} is the n_(T)×n_(T) channel matrix with independent and identically distributed (i.i.d.) entries, and

N={n_(t) ^(i)} is the T×n_(R) noise matrix with i.i.d. entires.

For an unknown channel matrix H, the conditional probability density function (pdf) of the received signal matrix Y given a transmitted signal matrix C is given by Formula 3:

$\begin{matrix} {{p\left( Y \middle| C \right)} = \frac{\exp\left( {{- {tr}}\left\{ {{Y^{\dagger}\left( {I + {\rho\;{CC}^{\dagger}}} \right)}^{- 1}Y} \right\}} \right)}{\pi^{n_{R}T}{\det\left( {I + {\rho\;{CC}^{\dagger}}} \right)}^{n_{R}}}} & \left\lbrack {{Formula}\mspace{14mu} 3} \right\rbrack \end{matrix}$

wherein,

tr represents trace, and

† represents conjugate transpose.

Now, the differential unitary space-time modulation (DUSTM) will be explained.

Let V_(k) denote the n_(T)×n_(T) data matrix at the kth block wherein only the DUSTM with T=n_(T) is considered. The data matrix V_(k) forms a unitary space-time constellation V with

${V \equiv {\left\{ {{\left. {V(l)} \middle| {V(l)} \right. = {V(1)}^{l}},{l = 0},1,2,\ldots\mspace{14mu},{L - 1}} \right\}\mspace{14mu}{wherein}}},{{V(1)} = {{diag}\left( {{\mathbb{e}}^{j\frac{2\pi\; u_{1}}{L}},{\mathbb{e}}^{j\frac{2\pi\; u_{2}}{L}},\ldots\mspace{14mu},{\mathbb{e}}^{j\frac{2\pi\; u_{n_{T}}}{L}}} \right)}},{and}$

L is the cardinality of V.

For the convenience of explanation, the unitary space-time constellation V is simply expressed as follows: (L; [u₁, u₂, . . . , u_(n) _(T) ])

Let C_(k) denote the n_(T)×n_(T) transmitted signal matrix at the kth block. This matrix C_(k), and the received signal matrix (Y_(k)) for C_(k) are given by Formulas 4 and 5, respectively: C _(k) =V _(k) C _(k−1) , C ₀ =I  [Formula 4] Y _(k)=√{square root over (ρ)}C _(k) H+N _(k)  [Formula 5]

wherein,

N_(k) is the noise matrix at the kth block, and

H is the channel matrix which is constant during two consecutive blocks.

Similarly to differential phase shift keying (DPSK), the DUSTM receiver estimates data matrix V_(k) by observing two consecutive received signal matrices

${{\overset{\_}{Y}}_{k}\overset{\Delta}{=}\left\lbrack {Y_{k - 1}^{T}\mspace{14mu}\vdots\mspace{14mu} Y_{k}^{T}} \right\rbrack^{T}},$ wherein T represents transpose. Two transmitted signal matrices that affect Y _(k) are

${\overset{\_}{C}}_{k} = {\left\lbrack {C_{k - 1}^{T}\mspace{14mu}\vdots\mspace{14mu} C_{k}^{T}} \right\rbrack^{T}.}$ From the Formula 3, the conditional pdf of Y _(k) is given by Formula 6:

$\begin{matrix} {{p\left( {\overset{\_}{Y}}_{k} \middle| {\overset{\_}{C}}_{k} \right)} = {\frac{\exp\left( {{- {tr}}\left\{ {{{\overset{\_}{Y}}_{k}^{\dagger}\left( {I - {{\frac{\rho}{1 + {2\;\rho}}\begin{bmatrix} I \\ \ldots \\ V_{k} \end{bmatrix}}\left\lbrack {I\mspace{20mu}\vdots\mspace{14mu} V_{k}^{\dagger}} \right\rbrack}} \right)}{\overset{\_}{Y}}_{k}} \right\}} \right)}{{\pi^{2n_{T}n_{R}}\left( {1 + {2\rho}} \right)}^{n_{T}n_{R}}}\mspace{110mu} = {p\left( {\overset{\_}{Y}}_{k} \middle| V_{k} \right)}}} & \left\lbrack {{Formula}\mspace{14mu} 6} \right\rbrack \end{matrix}$

The data matrix may be estimated by maximizing the conditional pdf of the Formula 6 as following Formula 7:

$\begin{matrix} {{\hat{V}}_{k} = {{\arg\mspace{11mu}{\max\limits_{V \in \;\nu}\mspace{11mu}{{tr}\left\{ {{{{\overset{\_}{Y}}_{k}^{\dagger}\begin{bmatrix} I \\ \ldots \\ V \end{bmatrix}}\left\lbrack {I\mspace{20mu}\vdots\mspace{14mu} V^{\dagger}} \right\rbrack}{\overset{\_}{Y}}_{k}} \right\}}}}\mspace{25mu} = {\arg\mspace{11mu}{\max\limits_{V \in \nu}{{Re}\mspace{11mu}{tr}\left\{ {Y_{k - 1}Y_{k}^{\dagger}V} \right\}}}}}} & \left\lbrack {{Formula}\mspace{14mu} 7} \right\rbrack \end{matrix}$

Hereinafter, the method of differential unitary space-time modulation by encoding messages to transmit through multiple transmit antenna into multiple trellis code (MTC).

FIG. 1 illustrates an exemplary transmitter used in the MTC DUSTM according to the present invention. Information bits are encoded by a rate mb/mn multiple trellis code. The encoded bit stream is divided into the blocks of n bits each of which is mapped into an element in the unitary space-time constellation V. After matrix-wise interleaving, the unitary space-time mapped signals are differentially encoded and transmitted by transmit antennas over a slow flat Rayleigh fading channel.

Let Y and V denote the received signal matrix sequence {Y₀, Y₁, . . . }, and the coded signal matrix sequence {V₀, V₁, . . . }, respectively. The pair-wise error probability (PEP), i.e., the probability of incorrectly decoding V to U, is given by Formula 8:

$\begin{matrix} {{p\left( V\rightarrow U \right)} \leq {\prod\limits_{k}{\det\left( {I + {\frac{\rho^{2}}{4\left( {1 + {2\rho}} \right)}\left( {V_{k} - U_{k}} \right)^{\dagger}\left( {V_{k} - U_{k}} \right)}} \right)}^{- n_{R}}}} & \left\lbrack {{Formula}\mspace{14mu} 8} \right\rbrack \end{matrix}$

Let η (V,U) denote the set of k such that V_(k)

U_(k). Then inequality of the Formula 8 becomes Formula 9:

$\begin{matrix} {{p\left( V\rightarrow U \right)} \leq {\left( \frac{\rho}{8} \right)^{{- n_{T}}n_{R}\delta}{\prod\limits_{k \in \eta}{{\det\left( {V_{k} - U_{k}} \right)}}^{{- 2}n_{R}}}}} & \left\lbrack {{Formula}\mspace{14mu} 9} \right\rbrack \end{matrix}$

wherein, δ is the size of η (V,U), i.e., the block Hamming distance between V and U.

To minimize the PEP of the Formula 9, it is needed to maximize the minimum block Hamming distance

$\left( {\delta_{\min} \equiv {\min\limits_{U}{{\eta\left( {V,U} \right)}}}} \right),$ and the minimum product of squared determinant distance ((ΠD²)_(min)). the minimum product of squared determinant, (ΠD²)_(min) is defined as following Formula 10:

$\begin{matrix} {{\left( {\prod D^{2}} \right)_{\min} \equiv {\min\limits_{U}\left( {\prod\limits_{k \in {\eta{({V,U})}}}{D^{2}\left( {V_{k},U_{k}} \right)}} \right)}}\mspace{121mu} = {\min\limits_{U}\left( {\prod\limits_{k \in {\eta{({V,U})}}}{{\det\left( {V_{k} - U_{k}} \right)}}^{\frac{2}{n_{T}}}} \right)}} & \left\lbrack {{Formula}\mspace{14mu} 10} \right\rbrack \end{matrix}$

Let A₀ denote the complete constellation, i.e., signal points 0, 1, 2, . . . , L−1, and A₀{circle around (×)}A₀ denote a two-fold ordered Cartesian product of A₀ with itself wherein the ordered Cartesian product means the concatenation of corresponding elements in the two sets forming the product.

The partitioning is as follows:

The first step is to partition A₀{circle around (×)}A₀ into L signal sets A₀{circle around (×)}B_(i) (i=0, 1, . . . , L−1), wherein the jth element (j=0, 1, . . . , L−1) of B_(i) is nj⊕i, and ⊕ is modulo L.

The jth 2-tuple signal points from A₀{circle around (×)}B_(i) are the ordered pair (j,nj⊕i).

Then, n is chosen to maximize the minimum product of squared distance (ΠD²)_(min) with δ_(min)=m within the partitioned set. Let n* denote optimum n satisfying this condition. Then, the n* for δ_(min)=2 is given as following Formula 11. The n* is 1,3 for L=8, and 3,5 for L=16 from the Formula 11:

$\begin{matrix} {n^{*} = {{\arg\mspace{11mu}{\max\limits_{n}{\prod D^{2}}}}\mspace{25mu} = {\arg\;{\max\limits_{{n = 1},3,\ldots\;,{{L/2} - 1}}\left\{ {\min\limits_{{m = 1},2,\ldots\;,{{L/2} - 1}}{16\;{\sin\left( \frac{{mu}_{1}\pi}{L} \right)}\mspace{56mu}{\sin\left( \frac{{nmu}_{1}\pi}{L} \right)}{\sin\left( \frac{{mu}_{2}\pi}{L} \right)}{\sin\left( \frac{{nmu}_{2}\pi}{L} \right)}}} \right\}}}}} & \left\lbrack {{Formula}\mspace{14mu} 11} \right\rbrack \end{matrix}$

The second step is an odd-even split of the first level partitioning. The third and succeeding steps are identical to the second step, odd-even split.

FIG. 2 illustrates the set partitioning for (8; [1,3]) for multiple (m=2) trellis code with n*=3. At the first level of the set partitioning, the sets partitioned from the set A₀{circle around (×)}A₀ are given as follow:

${{A_{0} \otimes B_{0}} = \begin{bmatrix} 0 & 0 \\ 1 & 3 \\ 2 & 6 \\ 3 & 1 \\ 4 & 4 \\ 5 & 7 \\ 6 & 2 \\ 7 & 5 \end{bmatrix}},{{A_{0} \otimes B_{2}} = \begin{bmatrix} 0 & 2 \\ 1 & 5 \\ 2 & 0 \\ 3 & 3 \\ 4 & 6 \\ 5 & 1 \\ 6 & 4 \\ 7 & 7 \end{bmatrix}},{{A_{0} \otimes B_{4}} = \begin{bmatrix} 0 & 4 \\ 1 & 7 \\ 2 & 2 \\ 3 & 5 \\ 4 & 0 \\ 5 & 3 \\ 6 & 6 \\ 7 & 1 \end{bmatrix}},{{A_{0} \otimes B_{6}} = \begin{bmatrix} 0 & 6 \\ 1 & 1 \\ 2 & 4 \\ 3 & 7 \\ 4 & 2 \\ 5 & 5 \\ 6 & 0 \\ 7 & 3 \end{bmatrix}},$

Since the sets A₀{circle around (×)}B_(i) (i=0, 2, 4, 6) have the largest ΠD² in A₀{circle around (×)}A₀, sets A₀{circle around (×)}B_(i) (i=1, 3, 5, 7) are excluded from the set partitioning. If higher data rate is required, sets A₀{circle around (×)}B_(i) (i=1, 3, 5, 7) may be included in the set partitioning with sacrificing good distance property. The sets of the second level are odd-even split of the first level partitioning, and given as follows:

$\begin{matrix} {{{C_{0} \otimes D_{00}} = \begin{bmatrix} 00 \\ 26 \\ 44 \\ 62 \end{bmatrix}},{{C_{1} \otimes D_{01}} = \begin{bmatrix} \begin{matrix} 13 \\ 31 \end{matrix} \\ 57 \\ 75 \end{bmatrix}},{{C_{0} \otimes D_{20}} = \begin{bmatrix} 02 \\ 20 \\ 46 \\ 64 \end{bmatrix}},{{C_{1} \otimes D_{21}} = \begin{bmatrix} 15 \\ 33 \\ 51 \\ 77 \end{bmatrix}},} \\ {{{C_{0} \otimes D_{40}} = \begin{bmatrix} 04 \\ 22 \\ 40 \\ 66 \end{bmatrix}},{{C_{1} \otimes D_{41}} = \begin{bmatrix} \begin{matrix} 17 \\ 35 \end{matrix} \\ 53 \\ 71 \end{bmatrix}},{{C_{0} \otimes D_{60}} = \begin{bmatrix} 06 \\ 24 \\ 42 \\ 60 \end{bmatrix}},{{C_{1} \otimes D_{61}} = \begin{bmatrix} 11 \\ 37 \\ 55 \\ 73 \end{bmatrix}}} \end{matrix}$

Similarly, from the sets of the second level, the sets of the third level are given as follows:

$\begin{matrix} {{{E_{0} \otimes F_{00}} = \begin{bmatrix} 00 \\ 44 \end{bmatrix}},{{E_{1} \otimes F_{26}} = \begin{bmatrix} 26 \\ 62 \end{bmatrix}},{{E_{0} \otimes F_{13}} = \begin{bmatrix} 13 \\ 57 \end{bmatrix}},{{E_{1} \otimes F_{31}} = \begin{bmatrix} 31 \\ 75 \end{bmatrix}},} \\ {{{E_{0} \otimes F_{02}} = \begin{bmatrix} 02 \\ 46 \end{bmatrix}},{{E_{1} \otimes F_{20}} = \begin{bmatrix} 20 \\ 64 \end{bmatrix}},{{E_{0} \otimes F_{15}} = \begin{bmatrix} 15 \\ 51 \end{bmatrix}},{{E_{1} \otimes F_{33}} = \begin{bmatrix} 33 \\ 77 \end{bmatrix}},} \\ {{{E_{0} \otimes F_{04}} = \begin{bmatrix} 04 \\ 40 \end{bmatrix}},{{E_{1} \otimes F_{22}} = \begin{bmatrix} 22 \\ 66 \end{bmatrix}},{{E_{0} \otimes F_{17}} = \begin{bmatrix} 17 \\ 53 \end{bmatrix}},{{E_{1} \otimes F_{15}} = \begin{bmatrix} 35 \\ 71 \end{bmatrix}},} \\ {{{E_{0} \otimes F_{06}} = \begin{bmatrix} 06 \\ 42 \end{bmatrix}},{{E_{1} \otimes F_{24}} = \begin{bmatrix} 24 \\ 60 \end{bmatrix}},{{E_{0} \otimes F_{11}} = \begin{bmatrix} 11 \\ 55 \end{bmatrix}},{{E_{1} \otimes F_{37}} = \begin{bmatrix} 37 \\ 73 \end{bmatrix}}} \end{matrix}$

Based on the set partitioning described above, the multiple trellis codes are constructed as follows.

All elements in the constellation should be equally probable. The 2 branches departing from or converging to a state are assigned with elements from one of {A₀{circle around (×)}B₀, A₀{circle around (×)}B₄} and {A₀{circle around (×)}B₂, A₀{circle around (×)}B₆}. The branches departing from or converging to adjacent states are assigned with elements from the other.

While minimum product of squared determinant distance ΠD² in the sets of the first level is equal to each other, δ_(min) having the ΠD² in {A₀{circle around (×)}B₀, A₀{circle around (×)}B₄} and {A₀{circle around (×)}B₂, A₀{circle around (×)}B₆} is larger then that in {A₀{circle around (×)}B₀, A₀{circle around (×)}B₂} and {A₀{circle around (×)}B₄, A₀{circle around (×)}B₆}.

Since the error path between the parallel branches becomes the shortest error event path whose Hamming distance is δ_(min), parallel branches are assigned with elements from the set having the largest product of squared determinant distance at the lowest level of set partitioning. Hence the multiplicity of multiple trellis code guarantees the same δ_(min) as that of trellis code without parallel branches.

FIGS. 3 a and 3 b illustrate the multiple trellis code for (8; [1,3]). They show the rate 4/6 4-state, and 8-state multiple trellis code for (8; [1,3]), respectively. The values of ΠD² for the multiple trellis coded DUSTM for 4-state and 8-state are 4 and 16, respectively. The value of δ_(min) is 2.

Rate 6/8 multiple trellis codes for (16; [1,7]) are constructed by the same method by which multiple trellis for (8; [1,3]) are constructed.

FIGS. 4 a and 4 b illustrate bit error rate (BER) of multiple trellis coded (MTC) DUSTM compared to that of the uncoded DUSTM and trellis coded DUSTM for 2 transmit antenna and 1 receive antenna.

FIG. 4 a shows the BER of the MTC DUSTM for spectral efficiency R=1 bit/s/Hz. In FIG. 4 a, it is shown that the 4-state and 8-state MTC DUSTM give 0.9 dB and 1.0 dB gain over the 4-state and the 8-state trellis coded DUSTM at a BER of 10⁻⁵, respectively.

FIG. 4 b shows the BER of the MTC DUSTM for spectral efficiency R=1.5 bits/s/Hz. In FIG. 4 b, it is shown that the 8-state and 16-state MTC DUSTM give 2.5 dB and 0.7 dB gain over the 8-state and 16-state trellis coded DUSTM at a BER of 10⁻⁵, respectively.

Hereinafter, the computational complexity will be explained.

Since the number of branches per a transition in the trellis increases exponentially as spectral efficiency increases in a multiple trellis code, it has much higher computational complexity than a trellis code.

The computational complexity may be decreased by reducing the number of branches as follows.

First, the branch metrics for all elements in the unitary space-time constellation V at each block is computed and stored. Then, most-likely candidates within each set assigned to the parallel branches are found, and maximum likelihood decision among only the most-likely candidates is performed. The sets assigned to the parallel branches are the ordered Cartesian product of V. Once the branch metrics for the elements of each set are computed and stored, the branch metrics for repeated elements need not to be calculated again. Viterbi algorithm is performed with such selected branches.

The computational complexity is evaluated in terms of the number of operations such as addition, multiplication, and comparison. Let Mb denote the number of operations needed to compute a branch metric. For example, M_(b)=10 for (8; [1,3]), i.e., 8 multiplications and 2 additions are performed. Let M_(t) denote the number of operations needed to decode a symbol. Then, the number of operations in the MTC DUSTM is given by following Formula (12):

$\begin{matrix} {M_{t} = \frac{{2^{b + 1} \times m \times M_{b}} + {2 \times \left( {2^{{mb} + 1} + {2^{b} \times S}} \right)}}{m \times n_{T}}} & {\left\lbrack {{Formula}\mspace{25mu} 12} \right\rbrack\mspace{14mu}} \end{matrix}$

wherein S is the number of states.

The number of operations in the trellis coded DUSTM is given by Formula 13:

$\begin{matrix} {M_{t} = \frac{{2^{b + 1} \times M_{b}} + {2 \times 2^{b} \times S}}{n_{T}}} & {\left\lbrack {{Formula}\mspace{25mu} 13} \right\rbrack\mspace{14mu}} \end{matrix}$

Table 1 shows the number of operations in the MTC DUSTM compared to that in the trellis coded DUSTM. It is shown that the number of operations in the former is comparable to that in the latter.

TABLE 1 Number of operations in the MTC DUSTM R bits/s/Hz Type of coded DUSTM M_(t) 1  4-state MTC(8; [1, 3]) 64  8-state MTC (8; [1, 3]) 72  4-state trellis coded (8; [1, 3]) 56  8-state trellis coded (8; [1, 3]) 72 1.5  8-state MTC (16; [1, 7]) 176 16-state MTC (16; [1, 7]) 208  8-state trellis coded (16; [1, 7]) 114 16-state trellis coded (16; [1, 7]) 208

FIG. 5 is a flow chart illustrating the method of DUSTM by encoding messages to transmit through n_(T) transmit antennas in a communication system wherein the signals are transmitted between n_(T) transmit antennas and n_(R) receive antennas wherein n_(T) is an integer no less than 2, and n_(R) is an integer no less than 1.

The first step (S10) is to set-partition two-fold ordered Cartesian product of A₀, A₀{circle around (×)}A₀ into L signal sets A₀{circle around (×)}B_(i) (i=0, 1, . . . , L−1) The jth element (j=0, 1, . . . , L−1) of B_(i) is njδi, ⊕ is modulo L, A₀ is the complete constellation (signal point 0, 1, 2, . . . , L−1), the jth 2-tuple signal points is (j,nj⊕i) of the A₀{circle around (×)}B_(i), and n is selected from n* which maximize the minimum product of squared distance (ΠD²)_(min) with δ_(min)=m within the partitioned set using the Formula (11).

The second step (S20) is odd-even splitting the set partition. The odd-even split is repeated to obtain a multiple trellis code in a n_(T)×n_(T) matrix (S30).

Then, messages are encoded by the multiple trellis code (S40), and the encoded messages are modulated by a unitary space-time modulation method according to Formula 2 (S50). The relation among the transmitted signal C from the jth transmit antenna (j=1, 2, . . . , n_(T)) at time t (t=1, 2, . . . , T), the received signal Y at the ith receive antenna for the transmitted signal C, channel matrix H, and noise matrix N is given by the Formula 2.

The foregoing embodiments of the present invention are merely exemplary and are not to be construed as limiting the present invention. Many alternatives, modifications and variations will be apparent to those skilled in the art.

As described above, the modulation method according to the present invention shows better performance. The MTC DUSTM achieves smaller BER than the trellis coded DUSTM without increase of the computational complexity for the same spectral efficiency. 

1. A method of differential unitary space-time modulation by encoding messages to transmit through n_(T) transmit antennas in a communication system wherein the signals are transmitted between n_(T) transmit antennas and n_(R) receive antennas wherein n_(T) is an integer no less than 2, and n_(R) is an integer no less than 1, comprising the step of: encoding a message into multiple trellis code which maximizes a product of minimum Euclidian distances between signals having minimum Hamming distance, wherein a relation among a transmitted signal C from a jth transmit antenna (j=1, 2, . . . , n_(T)) at time t (t=1, 2, . . . , T), the received signal Y at the ith receive antenna for the transmitted signal C, channel matrix H, and noise matrix N is given by Formula 2: Y=√{square root over (ρ)}CH+N  [Formula 2] wherein, Y={Y_(t) ^(i)} is the n_(T×n) _(T) received signal matrix, C={C_(tj)} is the n_(T×n) _(R) transmitted signal matrix, H={h_(j) ^(i)} is the n_(T)×n_(T) channel matrix, N={n_(t) ^(i)} is the n_(T×n) _(R) noise matrix, ρ is the SNR of the receive antenna, and T is the symbolic period, wherein the symbolic period T is n_(T); a unitarty space time constellation V for V_(k), which is an n_(T)×n_(T) data matrix at the kth block among T symbolic period, is V ≡ {V(l)❘V(l) = V(l)^(l), l = 0, 1, 2, …  , L − 1} ${wherein},{{V(1)} = {{diag}\left( {{\mathbb{e}}^{j\frac{2\pi\; u_{1}}{L}},{\mathbb{e}}^{j\frac{2\pi\; u_{2}}{L}},\ldots\mspace{14mu},{\mathbb{e}}^{j\frac{2\pi\; u_{n_{T}}}{L}}} \right)}},{and}$ L is the cardinality of V; and the n_(T)×n_(T) transmitted signal matrix C_(k) at the kth block, and the received signal matrix Y_(k) for the C_(k) are given by Formulas 4 and 5, respectively: C _(k) =V _(k) C _(k−1) , C ₀ =I  [Formula 4] Y _(k) =√{square root over (ρ)}C _(k) H+N _(k)wherein,  [Formula 5] wherein, N_(k) is the noise matrix at the kth block, and H is the channel matrix which is constant during two consecutive blocks; and wherein pair-wise error probability (PEP) which is the probability of incorrectly decoding V to U when the encoded signal matrix sequence V={V₀, V₁, . . . } is transmitted, is given by Formula 9, and in order to minimize the PEP of the Formula (9), the message is encoded to maximize the minimum block Hamming distance δ_(min) defined as Formula 14, and the minimum product of squared determinant distance ((ΠD²)_(min)) defined as Formula 10, wherein: $\begin{matrix} {{{p\left( V\rightarrow U \right)} \leq {\left( \frac{\rho}{8} \right)^{{- n_{T}}n_{R}\delta}{\prod\limits_{k \in \eta}\;{{\det\left( {V_{k} - U_{k}} \right)}}^{{- 2}n_{R}}}}}\;} & \left\lbrack {{Formula}\mspace{20mu} 9} \right\rbrack \\ {{\left( {\prod\; D^{2}} \right)_{\min} \equiv {\min\limits_{U}\left( {\prod\limits_{k \in {\eta{({V,U})}}}\;{D^{2}\left( {V_{k},U_{k}} \right)}} \right)}} = {\min\limits_{U}\left( {\prod\limits_{k \in {\eta{({V,U})}}}{{\det\left( {V_{k} - U_{k}} \right)}}^{\frac{2}{n_{T}}}} \right)}} & \left\lbrack {{Formula}\mspace{14mu} 10} \right\rbrack \\ {\delta_{\min} \equiv {\min\limits_{U}{{\eta\left( {V,U} \right)}}}} & \left\lbrack {{Formula}\mspace{20mu} 14} \right\rbrack \end{matrix}$ wherein, δ is the size of η(V,U) and represents the block Hamming distance between V and U, and η(V,U) is the set of k such that V_(k)≠U_(k).
 2. The method according to claim 1, wherein a partitioning process for encoding the message comprises steps of: partitioning two-fold ordered Cartesian product of A₀, A₀{circle around (×)}A₀ into L signal sets A₀{circle around (×)}B_(i) (i=0, 1, . . . , L−1), wherein the jth element (j=0, 1, . . . , L−1) of B_(i) is nj⊕i, ⊕ is is modulo L, and A₀ is the complete constellation (signal point 0, 1, 2, . . . , L−1); obtaining the jth 2-tuple signal points (j,nj⊕i) from the A₀{circle around (×)}B_(i) choosing n* which maximizes the minimum product of squared distance (ΠD²)_(min) with δ_(min)=m within the partitioned set using the Formula (11); and odd-even splitting the set partition and repeating the odd-even split: $\begin{matrix} {n^{*} = {{\arg{\max\limits_{n}{\prod D^{2}}}} = {\arg{\max\limits_{{n = 1},3,\;\ldots\;,{{L/2} - 1}}{\left\{ {\min\limits_{{m = 1},2,\;\ldots\;,{{L/2} - 1}}{16{\sin\left( \frac{{mu}_{1}\pi}{L} \right)}{\sin\left( \frac{{nmu}_{1}\pi}{L} \right)}{\sin\left( \frac{{mu}_{2}\pi}{L} \right)}{\sin\left( \frac{{nmu}_{2}\pi}{L} \right)}}} \right\}.}}}}} & \left\lbrack {{Formula}\mspace{20mu} 11} \right\rbrack \end{matrix}$
 3. A method of differential unitary space-time modulation by encoding messages to transmit through n_(T) transmit antennas in a communication system wherein the signals are transmitted between n_(T) transmit antennas and n_(R) receive antennas wherein n_(T) is an integer no less than 2, and n_(R) is an integer no less than 1, the method comprising the steps of: (a) set-partitioning two-fold ordered Cartesian product of A₀, A₀{circle around (×)}A₀ into L signal sets A₀{circle around (×)}B_(i) (i=0, 1, . . . , L−1), wherein the jth element (j=0, 1, . . . , L−1) of B_(i) is nj⊕i, ⊕ is modulo L, A₀ is the complete constellation (signal point 0, 1, 2, . . . , L−1), the jth 2-tuple signal points is (j,nj⊕i) of the A₀{circle around (×)}B_(i), and n is selected from n* which maximize the minimum product of squared distance (ΠD²)_(min) with δ_(min)=m within the partitioned set using the Formula (11); (b) odd-even splitting the set partition and repeating the odd-even split to obtain a multiple trellis code in a n_(T)×n_(T) matrix; (c) encoding messages by the multiple trellis code; and (d) modulating the encoded message by a unitary space-time modulation method according to Formula 2, wherein the relation among the transmitted signal C from the jth transmit antenna (j=1, 2, . . . , n_(T)) at time t (t=1, 2, . . . , T), the received signal Y at the ith receive antenna for the transmitted signal C, channel matrix H, and noise matrix N is given by Formula 2: Y=√{square root over (ρ)}CH+N  [Formula 2] wherein, Y={Y_(t) ^(i)} is the n_(T)×n_(T) received signal matrix, C={C_(tj)} is the n_(T)×n_(R) transmitted signal matrix, H={h_(j) ^(i)} is the n_(T)×n_(T) channel matrix, N={n_(t) ^(i)} is the n_(T)×n_(R) noise matrix, ρ is the SNR of the receive antenna, and T is the symbolic period, $\begin{matrix} {n^{*} = {{\arg{\max\limits_{n}{\prod D^{2}}}} = {\arg{\max\limits_{{n = 1},3,\;\ldots\;,{{L/2} - 1}}{\left\{ {\min\limits_{{m = 1},2,\;\ldots\;,{{L/2} - 1}}{16{\sin\left( \frac{{mu}_{1}\pi}{L} \right)}{\sin\left( \frac{{nmu}_{1}\pi}{L} \right)}{\sin\left( \frac{{mu}_{2}\pi}{L} \right)}{\sin\left( \frac{{nmu}_{2}\pi}{L} \right)}}} \right\}.}}}}} & \left\lbrack {{Formula}\mspace{20mu} 11} \right\rbrack \end{matrix}$ 